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Cover
Preface
1 The Mystery of Wave Propagation and Radiation from an Antenna
1. Summary
2. 1.1 Historical Overview of Maxwell’s Equations
3. 1.2 Review of Maxwell–Hertz–Heaviside Equations
4. 1.3 Development of Wave Equations
5. 1.4 Methodologies for the Solution of the Wave Equations
6. 1.5 General Solution of Maxwell’s Equations
7. 1.6 Power (Correlation) Versus Reciprocity (Convolution)
8. 1.7 Radiation and Reception Properties of a Point Source Antenna in Frequency and in Time Domain
9. 1.8 Radiation and Reception Properties of Finite‐Sized Dipole‐Like Structures in Frequency and in Time
10. 1.9 An Expose on Channel Capacity
11. 1.10 Conclusion
12. References
2 Characterization of Radiating Elements Using Electromagnetic Principles in the Frequency Domain
1. Summary
2. 2.1 Field Produced by a Hertzian Dipole
3. 2.2 Concept of Near and Far Fields
4. 2.3 Field Radiated by a Small Circular Loop
5. 2.4 Field Produced by a Finite‐Sized Dipole
6. 2.5 Radiation Field from a Finite‐Sized Dipole Antenna
7. 2.6 Maximum Power Transfer and Efficiency
8. 2.7 Radiation Efficiency of Electrically Small Versus Electrically Large Antenna
9. 2.8 Challenges in Designing a Matched ESA
10. 2.9 Near‐ and Far‐Field Properties of Antennas Deployed Over Earth
11. 2.10 Use of Spatial Antenna Diversity
12. 2.11 Performance of Antennas Operating Over Ground
13. 2.12 Fields Inside a Dielectric Room and a Conducting Box
14. 2.13 The Mathematics and Physics of an Antenna Array
15. 2.14 Does Use of Multiple Antennas Makes Sense?
16. 2.15 Signal Enhancement Methodology Through Adaptivity on Transmit Instead of MIMO
17. 2.16 Conclusion
18. Appendix 2A Where Does the Far Field of an Antenna Really Starts Under Different Environments?
19. References
3 Mechanism of Wireless Propagation
1. Summary
2. 3.1 Introduction
3. 3.2 Description and Analysis of Measured Data on Propagation Available in the Literature
4. 3.3 Electromagnetic Analysis of Propagation Path Loss Using a Macro Model
5. 3.4 Accurate Numerical Evaluation of the Fields Near an Earth–Air Interface
6. 3.5 Use of the Numerically Accurate Macro Model for Analysis of Okumura et al.’s Measurement Data
7. 3.6 Visualization of the Propagation Mechanism
8. 3.7 A Note on the Conventional Propagation Models
9. 3.8 Refinement of the Macro Model to Take Transmitting Antenna’s Electronic and Mechanical Tilt into Account
10. 3.9 Refinement of the Data Collection Mechanism and its Interpretation Through the Definition of the Proper Route
11. 3.10 Lessons Learnt: Possible Elimination of Slow Fading and a Better Way to Deploy Base Station Antennas
12. 3.11 Cellular Wireless Propagation Occurs Through the Zenneck Wave and not Surface Waves
13. 3.12 Conclusion
14. Appendix 3A Sommerfeld Formulation for a Vertical Electric Dipole Radiating Over an Imperfect Ground Plane
15. Appendix 3B Asymptotic Evaluation of the Integrals by the Method of Steepest Descent
16. Appendix 3C Asymptotic Evaluation of the Integrals When there Exists a Pole Near the Saddle Point
17. Appendix 3D Evaluation of Fields Near the Interface
18. Appendix 3E Properties of a Zenneck Wave
19. Appendix 3F Properties of a Surface Wave
20. References
4 Methodologies for Ultrawideband Distortionless Transmission/Reception of Power and Information
1. Summary
2. 4.1 Introduction
3. 4.2 Transient Responses from Differently Sized Dipoles
4. 4.3 A Travelling Wave Antenna
5. 4.4 UWB Input Pulse Exciting a Dipole of Different Lengths
6. 4.5 Time Domain Responses of Some Special Antennas
7. 4.6 Two Ultrawideband Antennas of Century Bandwidth
8. 4.7 Experimental Verification of Distortionless Transmission of Ultrawideband Signals
9. 4.8 Distortionless Transmission and Reception of Ultrawideband Signals Fitting the FCC Mask
10. 4.9 Simultaneous Transmission of Information and Power in Wireless Antennas
11. 4.10 Effect of Broadband Matching in Simultaneous Information and Power Transfer
12. 4.11 Conclusion
13. References
Index
End User License Agreement

List of Tables

Chapter 02
1. Table 2.1 The simulation results for all the previous cases A, B, and C.
2. Table 2.2 The simulation results for the unmatched 3λ/4 dipole and the unmatched 0.65λ dipole cases.
3. Table 2.3 Received power by a 50 Ω load connected to dipole antenna of various lengths.
4. Table 2.4 Received power by a load, conjugately matched to the input impedance of the antenna, connected to dipole antenna with various lengths.
5. Table 2.5 Transfer of power between a transmitting and a receiving antenna for a fixed input power for various heights and separation distances.
6. Table 2.6 Transfer of power between a transmitting horn and a receiving dipole antenna for a fixed input power for various heights and separation distances.
7. Table 2.7 Transfer of power between a transmitting dipole and a receiving dipole antenna for a fixed input power and for two different scenarios (one inside a box and the other operating in free space).
8. Table 2.8 Input impedance of the half wave dipole antenna when operating inside a perfectly conducting box of different sizes.
9. Table 2.9 Input impedance of the half wave dipole antenna when operating inside a dielectric box of thickness of 0.1 λ with different dimensions and having a relative permittivity of ε_r = 2 ─ j 2 and a relative permeability of μ_r = 2 ─ j 2.
10. Table 2.10 Input impedance of the half wave dipole antenna when operating inside a dielectric box of thickness of 0.1 λ with different dimensions and having a relative permittivity of ε_r = 10 ─ j 10 and a relative permeability of μ_r = 10 ─ j 10.
11. Table 2.11 Magnitude of the currents measured at the three receiving antennas 1, 2, and 3 for three different choices of excitations.
12. Table 2.12 Complex values of the currents measured at the three antennas located on the three receivers marked as 4, 5, and 6 for three different choices of excitations given by (2.78).
Chapter 03
1. Table 3.1 Values for the received power obtained at a stationary half‐wave receiving dipole and radiated by a half wavelength vertical half‐wave dipole located at different radial distances and for different heights over the ground.
2. Table 3.2 Values for the received power obtained at a stationary half‐wave receiving dipole and radiated by a loop of half wavelength on each side located at different radial distances and for different heights over the ground.
Chapter 04
1. Table 4.1 The different weights used over each frequency band.
2. Table 4.2 The relationship between VSWR and the reflected power.
3. Table 4.3 Frequency range in use in the presence of VSWR constraint.

List of Illustrations

Chapter 01
1. Figure 1.1 A general solution of the wave equation.
2. Figure 1.2 Propagation of an electromagnetic wave in free space.
3. Figure 1.3 Temporal and spatial variation of a propagating electromagnetic wave in free space.
4. Figure 1.4 Illustration of the principle of superposition for a purely resistive circuit.
5. Figure 1.5 Regions of response for a current patch with a cosine distribution to rectangular pulses of sinusoidal carrier for
6. Figure 1.6 Sinusoidally modulated pulse.
7. Figure 1.7 Far field along the elevation angle θ = 30° for case A.
8. Figure 1.8 Sinusoidally modulated short pulse.
9. Figure 1.9 Far field along the elevation angle θ = 30° for case B.
10. Figure 1.10 Rectangular pulse.
11. Figure 1.11 Far field along the elevation angle θ = 30^∘ for case A.
12. Figure 1.12 Far field along the elevation angle θ = 30^° for case B.
Chapter 02
1. Figure 2.1 An equivalent circuit of a Hertzian dipole.
2. Figure 2.2 A loop made of very thin wire.
3. Figure 2.3 The geometry of a dipole antenna and the location of the field point. The curve on the left represents the current distribution on the wire antenna.
4. Figure 2.4 A simple circuit for a voltage divider. (a) Real value for the load and (b) Complex value for the load.
5. Figure 2.5 P_out/P_{out − max} versus R_L and X_L.
6. Figure 2.6 Efficiency versus R_L and X_L.
7. Figure 2.7 Efficiency versus power transfer.
8. Figure 2.8 Performance of Antennas under different matching conditions.
9. Figure 2.9 (a) The magnitude of the electric field in dB as a function of the horizontal distance from the antenna for all the three cases A, B, and C. (b) Zoom in of Figure 2.9(a).
10. Figure 2.10 The magnitude of the electric field in dB as a function of the horizontal distance from an unmatched 3λ/4 dipole, unmatched 0.65λ dipole, unmatched half wave dipole, and matched half wave dipole.
11. Figure 2.11 The magnitude of the z‐component of the electric field in dB for the various dipole lengths (unmatched scenario).
12. Figure 2.12 The magnitude of the z‐component of the electric field in dB for the various dipole lengths (matched scenario).
13. Figure 2.13 S₁₁ in dB of various lengths of dipoles matched using an external passive network.
14. Figure 2.14 S₁₁ magnitude versus frequency for the meander line folded monopole and the simple monopole.
15. Figure 2.15 The radiation pattern of a half wave dipole in free space (only one fourth is shown).
16. Figure 2.16 Radiation pattern of a half wave‐length dipole situated on top of a 20‐m tower located over a perfectly conducting ground (only one fourth is shown).
17. Figure 2.17 Radiation pattern of a dipole (a) normal, and (b) expended, situated on top of a 20 m tower located over a perfectly conducting ground.
18. Figure 2.18 Radiation pattern of a half wave‐length dipole situated on top of a 20‐m tower located over an imperfectly conducting ground (only one fourth shown).
19. Figure 2.19 Radiation pattern of a dipole over a real ground. (a) normal, and (b) expended, situated on top of a 20 m tower located over an imperfectly conducting ground.
20. Figure 2.20 Radiation pattern of a half wave dipole elements located in various environments.
21. Figure 2.21 Different transmitting horn configurations at a height of 20 m, 20 m tilted downwards at an angle of 11°, 10 m, 2 m, and 1 m tilted upwards by an angle of 1° above the ground plane. The receiving dipole is located at 2 m above the ground plane and at a horizontal distance of 100 m from the transmitter. The LOS component between the transmitter and the receiver is blocked by a dielectric wall.
22. Figure 2.22 Different transmitting dipole configurations at a height of 20 m, 20 m tilted downwards at an angle of 11°, 10 m, 2 m, and 1 m tilted upwards by an angle of 1° above the ground plane. The receiving dipole is located at 2 m above the ground plane and at a horizontal distance of 100 m from the transmitter. The receiving dipole is enclosed by a dielectric shell.
23. Figure 2.23 The three components of the fields E_r,E_θ,E_ϕ in (a) free space and (b) inside the room at x = − 3.75λ, z = − 3.75λ, as a function of y.
24. Figure 2.24 The three components of the fields E_r, E_θ, E_ϕ (a) free space and (b) inside the room at x = − 0.0198λ, z = − 3.75, as a function of y.
25. Figure 2.25 The three components of the fields E_r, E_θ, E_ϕ (a) free space and (b) inside the room at x = − 3.75λ, z = − 0.0198λ as a function of y.
26. Figure 2.26 The three components of the fields E_r, E_θ, E_ϕ in (a) free space and (b) inside the room at x = − 0.0198λ, z = − 0.0198λ as a function of y.
27. Figure 2.27 A half‐wave dipole antenna radiating in (a) in free space and (b) operating inside a conducting cube.
28. Figure 2.28 A phased array (one‐dimensional) of isotropic omnidirectional point radiators as antenna elements.
29. Figure 2.29 A phased array (one‐dimensional) of Hertzian dipoles as antenna elements.
30. Figure 2.30 A single radiating antenna connected to a transmitter.
31. Figure 2.31 Two radiating antennas connected in parallel to a transmitter.
32. Figure 2.32 A MIMO system model.
33. Figure 2.33 Uncoupled parallel decomposition of a MIMO channel.
34. Figure 2.34 A SISO system enclosed in conducting concentric cylinders.
35. Figure 2.35 A 2 × 2 MIMO system enclosed in a conducting concentric box.
36. Figure 2.36 A multiple‐user transmit/receive scenario.
37. Figure 2.37 Simulation situation when the conducting inner cylinder is replaced by a dielectric cube.
38. Figure 2.38 A six helical antenna transmit/receive system.
39. Figure 2.39 Magnitude of the measured currents at the three receivers with W₄.
40. Figure 2.40 Magnitude of the measured currents at the three receivers with W₅.
41. Figure 2.41 Magnitude of the measured currents at the three receivers with W₆.
42. Figure 2.42 A six helical antenna transmit/receive system inside a conducting rectangular cylinder.
43. Figure 2A.1 The total electric field magnitude as a function of the azimuthal angle ϕ at (a) r = 0.85 m, (b) r = 3 m, (c) r = 100 m.
44. Figure 2A.2 (a) The ratio of |E_θ/H_φ| for a 0.1λ z‐oriented dipole antenna as a function of the radial distance in meters for various azimuthal orientations in θ. (b) The ratio |E_r/E_θ| in dB for a 0.1 λ z‐oriented dipole antenna as a function of the radial distance in meters for various directions in θ.
45. Figure 2A.3 (a) The ratio of |E_θ/H_φ| for a 0.5λ z‐oriented dipole antenna as a function of the radial distance in meters for various directions in θ. (b) The ratio of |E_r/E_θ| for a 0.5λ z‐oriented dipole antenna as a function of the radial distance in meters for various directions in θ.
46. Figure 2A.4 (a) The ratio of |E_θ/H_φ| for a 5 λ z‐oriented dipole antenna as a function of the radial distance in meters for various directions in θ. (b) The ratio |E_r/E_θ| in dB for a 5 λ z‐oriented dipole antenna as a function of the radial distance in meters for various directions in θ.
47. Figure 2A.5 (a) The ratio of |E_θ/H_φ| for a 10 λ z‐oriented dipole antenna located at a height of 60 m above an urban ground. (b) The ratio |E_r/E_θ| in dB for a 10 λ z‐oriented dipole antenna located at a height of 60 m above an urban ground.
48. Figure 2A.6 (a) The ratio of |E_θ/H_φ| for a 5 λ z‐oriented dipole antenna at height of 30 m above an urban ground. (b) The ratio |E_r/E_θ| in dB for a 5 λ z‐oriented dipole antenna at height of 30 m above an urban ground.
49. Figure 2A.7 (a) The ratio of |E_θ/H_φ| for a λ/2 z‐oriented dipole antenna at height of 30 m above an urban ground. (b) The ratio |E_r/E_θ| in dB for a λ/2 z‐oriented dipole antenna at height of 30 m above an urban ground.
Chapter 03
1. Figure 3.1 Distance dependence of median field strength in urban area (E ∝ d⁻ⁿ)
2. Figure 3.2 Prediction from Ericsson in‐building path loss model. (Simon R. Saunders, Advances in mobile propagation prediction methods, Chapter 3 of Mobile Antenna Systems Handbook, Ed. Kyohei Fujimoto, Artech House, 2008.
3. Figure 3.3 Empirical model of macro cell propagation measurements performed at 900 MHz. The dots are actual experimental data points taken in a suburban area, whereas the solid line represents the best fit empirical model.
4. Figure 3.4 (a) Experimental arrangement for the measurements of the received field strength over water carried out by Burrows in Seneca Lake located in New York State. (b) Measurements of the relative field strength with distance over water.
5. Figure 3.5 Variation of path loss exponent with distance for OM‐1 base station (1800 MHz) located in a dense urban environment. Base station antenna height is 24 m.
6. Figure 3.6 Variation of path loss exponent with distance for UA base station (900 MHz) in a medium urban environment. Base station antenna height is 24 m.
7. Figure 3.7 Variation of path loss exponent with distance for ARUNIA 19 base station (900 MHz) in an industrial area. Base station antenna height is 50 m.
8. Figure 3.8 Opportunity Window: The best frequencies for mobile broadband are high enough so that the antenna can be made conveniently compact yet not so high that signals will fail to penetrate buildings [9]. That leaves a narrow range of frequencies reliable for use (red band).
9. Figure 3.9 One way attenuation through common building materials as a function of frequency. Observe the curve does not even start till 4.5 GHz.
10. Figure 3.10 Different transmitting and receiving configurations for LOS and non‐LOS scenarios.
11. Figure 3.11 Received power at the dipole for the LOS and non‐LOS scenario.
12. Figure 3.12 A vertical dipole over a horizontal imperfect ground plane.
13. Figure 3.13 Comparison between the experimental (Okumura et al.) and theoretical predictions (Schelkunoff and Sommerfeld formulations) computed through a macro model for predicting propagation path loss in an urban environment at (a) 453, (b) 922, and (c) 1920 MHz.
14. Figure 3.14 The variation of the magnitude of the E_z component of the electric field from a half‐wavelength dipole as a function of distance, at an operating frequency of 1 GHz. The height of the observation point was 2 m from an urban ground with a permittivity ε_r = 4 and a conductivity of σ = 2 × 10⁻⁴.
15. Figure 3.15 Magnitude of the z‐component of the electric field in dB radiated from a half‐wave vertical dipole antenna over different imperfect grounds, height of the antenna is 20 m, different curve belongs to different imperfect ground, the frequency of operation is 900 MHz.
16. Figure 3.16 Equivalent image sources that generate the desired radiation fields at the interface. The image starts at –h.
17. Figure 3.17 Elongated image of the sun over a rippled lake produced by the wind flow. (http://fineartamerica.com/featured/alexandria‐bay‐sunset‐steve‐ohlsen.html)
18. Figure 3.18 Waves on a wet ground due to the partial reflectivity of the surface. The figures give an impression on the physical propagation mechanism to be expected in cellular environments at frequencies where a wet earth represents a complex impedance surface.
19. Figure 3.19 The dashed red curve is the received power from a vertical half‐wave dipole at 20 m above an urban ground using the electromagnetic analysis code AWAS. The dashed black curve is the results from the Hata model for the same setup as used in the AWAS simulation. The blue curve is the results for the two‐ray model for the same setup as used in the AWAS simulation.
20. Figure 3.20 Xpol 45° tilted antenna model.
21. Figure 3.21 Magnitude of the theta component of the electric field in dB radiated from five half‐wave dipoles located at a height 10 m above an urban ground.
22. Figure 3.22 Magnitude of the theta component of the electric field in dB radiated from five half‐wave dipoles at a height of 20 m above an urban ground.
23. Figure 3.23 Horizontal distance in meters versus (a) route index and (b) measured power versus route index for Galewela, Sri Lanka, where the environment is considered to be rural.
24. Figure 3.24 Comparison between the drive‐test data taken from Galewela and the macro modeling‐based simulation results. (a) The measured data along the selected radial route and (b) The measured data along the selected radial route after averaging it at each spatial sample point.
25. Figure 3.25 Horizontal distance versus route index and measured power versus route index for Udubeddawa where the environment is considered to be rural.
26. Figure 3.26 Comparison between the drive‐test data taken from Udubeddawa and the macro modeling‐based simulation results. (a) The measured data along the selected radial route and (b) The measured data along the selected radial route after averaging it at each spatial sample point.
27. Figure 3.27 Plot of the far‐field pattern for a dipole at different tilt angles and located at different heights above earth. The bottom of the dipole is located from the ground at (a) 0.01, (b) 0.25, (c) 0.5, (d) 0.75, (e) 1.0, and (f) 2.0 m.
28. Figure 3.28 (a) Plot of the horizontal component of the electric field as a function of the height of the transmitting antenna above the ground and for a particular plot (the black line) the antenna is tilted upwards towards the sky and away from the ground, resulting in elimination of fading. (b) Plot of the vertical component of the electric field as a function of the height of the transmitting antenna above the ground, and for a particular plot (the black line) the antenna is tilted upwards towards the sky and away from the ground, resulting in elimination of fading.
29. Figure 3.29 Measurement set up for recording the signal strength as a function of the height of the transmitting antenna and also for its various tilts with the vertical axis for a fixed location of the receiver. Three different experimental setups are displayed in a single figure.
30. Figure 3.30 Measurement of the electric field strength as a function of the height of the base station antenna for two different frequencies. The receiver is located 1.5 m off the ground.
31. Figure 3.31 Measurement of the electric field strength for two different heights and angular tilt of the base station antenna at 800 MHz.
32. Figure 3.32 Measurement of the electric field strength for two different heights and angular tilt of the base station antenna at 2.4 GHz.
33. Figure 3.33 Brewster‐Zenneck (TM reflection coefficient ρ_TM = 0) and surface plasmon (TM reflection coefficient ρ_TM = ∞) cases. (a) Brewster–Zenneck wave, reflection coefficient is zero and (b) surface wave, reflection coefficient is infinity.
34. Figure 3.34 Plot of the transverse component of the reflected electric fields at ρ = 100 m from the source.
35. Figure 3A.1 The contour of integration along the real axis from 0 to ∞ in the complex λ‐plane.
36. Figure 3A.2 Actual location of the pole in the lower complex λ‐plane and the contour C₁ enclosing them.
37. Figure 3A.3 The complex β plane showing possible branch points, branch cuts, poles, and the path of steepest descent for an imperfect ground plane with the material parameter ε = ε^' (1 − j)² and |ε| > > 1 .
Chapter 04
1. Figure 4.1 A simple C‐R circuit.
2. Figure 4.2 Response of the circuit.
3. Figure 4.3 Response on transmit.
4. Figure 4.4 Response on receive.
5. Figure 4.5 Gaussian‐shaped pulse used either as a voltage source or as an incident wave. The unit of time is in light‐meter (lm).
6. Figure 4.6 Spectrum of the Gaussian pulse which is used as an input excitation. The vertical axis represents the amplitude of the spectrum.
7. Figure 4.7 Single dipole of length 1m and width 0.04 m.
8. Figure 4.8 Radiated electric far‐field along φ = 0° for different elevation angles for the incident Gaussian pulse.
9. Figure 4.9 Induced current at edge 41 due to a θ‐polarized incident Gaussian pulse arriving from φ = 0° and for different incident elevation angles.
10. Figure 4.10 Two dipoles separated by 1 m along the y‐direction.
11. Figure 4.11 Radiated field for E_θ along φ = 45° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
12. Figure 4.12 Radiated field for E_φ along φ = 45° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
13. Figure 4.13 Radiated field for E_θ along φ = 90° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
14. Figure 4.14 Radiated field for E_φ along φ = 90° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
15. Figure 4.15 Radiated field for E_θ along φ = 135° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
16. Figure 4.16 Radiated field for E_φ along φ = 135° for different elevation angles from the two dipoles when one of them is center fed with a Gaussian pulse of Figure 4.5.
17. Figure 4.17 Two dipoles separated by 1 m along the x‐direction.
18. Figure 4.18 Induced current at edge 41 for one of the dipoles due to a θ‐polarized incident Gaussian pulse arriving from φ = 0° for different elevation angles.
19. Figure 4.19 (a) A typical monocycle input. (b) The frequency spectrum of the input.
20. Figure 4.20 (a) Frequency spectrum with σ/τ < 1. (b) Frequency spectrum with σ/τ > 1.
21. Figure 4.21 Radiation from the antenna with σ/τ < 1.
22. Figure 4.22 Reception of a monocycle pulse by the antenna with σ/τ < 1.
23. Figure 4.23 Radiation from the antenna 1 with a 50 ohms load at the feed.
24. Figure 4.24 Radiation from the antenna 1 with a 500 ohms load at the feed.
25. Figure 4.25 Radiation from the antenna with σ/τ > 1.
26. Figure 4.26 Reception of a monocycle pulse by the antenna with σ/τ > 1.
27. Figure 4.27 Resistive loading on the antenna with σ/τ < 1.
28. Figure 4.28 Resistive loading on the antenna with σ/τ > 1.
29. Figure 4.29 Radiation from Antenna 1 (σ/τ < 1) with loading.
30. Figure 4.30 Reception of a monocycle pulse by Antenna 2 (σ/τ < 1) with loading.
31. Figure 4.31 Radiation from Antenna 2 (σ/τ > 1) with loading.
32. Figure 4.32 Reception of a monocycle pulse by Antenna 2 (σ/τ > 1) with loading.
33. Figure 4.33 Radiation efficiency of the loaded dipole Antenna 1.
34. Figure 4.34 Input required for a monocycle transmission by the antenna.
35. Figure 4.35 Incident pulse shape to generate a monocycle of current at reception.
36. Figure 4.36 Structure of a biconical antenna.
37. Figure 4.37 Frequency spectrum of the transmitted signal for different flare angles of the bicone.
38. Figure 4.38 Radiated field from the bi‐ conical antenna for different flare angles.
39. Figure 4.39 Reception of a monocycle by a biconical antenna for different flare angles.
40. Figure 4.40 Structure of the biconical antenna with end caps.
41. Figure 4.41 Radiation from a biconical antenna with end caps.
42. Figure 4.42 Reception of a monocycle pulse by a biconical antenna with end caps.
43. Figure 4.43 Resistive loading profile along the length of the biconical antenna.
44. Figure 4.44 Radiation from the bicone antenna.
45. Figure 4.45 Reception of a monocycle by the bicone antenna.
46. Figure 4.46 Input required for a monocycle transmission.
47. Figure 4.47 Incident field required for a monocycle of induced current.
48. Figure 4.48 Structure of a TEM horn.
49. Figure 4.49 Radiation from a TEM horn antenna.
50. Figure 4.50 Reception of a monocycle by a TEM horn antenna.
51. Figure 4.51 Structure of a tapered horn antenna.
52. Figure 4.52 Radiation from a tapered horn antenna.
53. Figure 4.53 Reception of a monocycle by a tapered horn antenna.
54. Figure 4.54 Structure of a non‐resonating horn antenna.
55. Figure 4.55 Resisting loading profile on the antenna.
56. Figure 4.56 Radiation from non‐resonating tapered horn antenna.
57. Figure 4.57 Reception of a monocycle by a non‐resonating tapered horn antenna.
58. Figure 4.58 Radiation efficiency of the loaded tapered horn antenna.
59. Figure 4.59 Input required for a monocycle transmission.
60. Figure 4.60 Incident waveform shape to produce a monocycle of induced current.
61. Figure 4.61 A bi‐blade century band‐width antenna.
62. Figure 4.62 Radiation from a bi‐blade antenna.
63. Figure 4.63 Reception of a monocycle by a bi‐blade antenna.
64. Figure 4.64 Transient input resistance of a bi‐blade antenna.
65. Figure 4.65 Plot of the input impedance of a bi‐blade century bandwidth antenna with a 2.5 VSWR circle surrounding the input S‐parameter data.
66. Figure 4.66 Details of feeding the bi‐blade century bandwidth antenna with a coaxial transmission line.
67. Figure 4.67 A cone‐blade antenna structure for generating circular polarization.
68. Figure 4.68 Radiation from a cone‐blade antenna.
69. Figure 4.69 Reception of a monocycle by a cone‐blade antenna.
70. Figure 4.70 Transient input resistance of a cone‐blade antenna.
71. Figure 4.71 Comparison of the radiated field and the first derivative of the induced current when operating as a receiver.
72. Figure 4.72 (a) An Impulse Radiating Antenna (IRA). (b) shows a model of the full IRA, with the reflector segmented using 16 subsections around the quarter of the circumference (n = 16) for carrying out an accurate electromagnetic simulation up to 20GHz. (c) side view of the IRA.
73. Figure 4.73 Radiation from an IRA.
74. Figure 4.74 Reception of a monocycle pulse by the IRA.
75. Figure 4.75 Transient input resistance of the IRA.
76. Figure 4.76 Magnified view of the transient input resistance of the IRA.
77. Figure 4.77 Miniature UWB Antenna Range. Antennas shown include a conical antenna, TEM Horn antenna, D*dot probe antenna, and a Vertical Monopole Antenna.
78. Figure 4.78 4 Volt, 9 ps risetime, step pulse used for UWB antenna testing. Measured by an HP 50 GHz, 9 ps risetime, sampling oscilloscope. Scales are 750 mV/div & 10 ps/div.
79. Figure 4.79 Transmission from a conical antenna to a TEM horn antenna over a 25 cm path.
80. Figure 4.80 Transmission from a conical antenna to a TEM horn antenna over a 5 m path.
81. Figure 4.81 Transmission between a pair of identical conical antennas.
82. Figure 4.82 Transmission between a pair of identical TEM horn antennas.
83. Figure 4.83 Transmission from conical antenna to D*dot antenna. The lower trace is the output from the receive antenna. The upper trace is the integral of the lower trace.
84. Figure 4.84 Transmission from a D*dot antenna to a TEM horn antenna.
85. Figure 4.85 Transmission between a pair of identical D*dot antennas.
86. Figure 4.86 Transmission from monopole antenna to the TEM horn.
87. Figure 4.87 Transmission from the conical to the monopole. Lower trace receiving antenna output. Upper trace is dV/dt.
88. Figure 4.88 Transmission between a pair of identical 10 cm monopole antennas (25 ps/div).
89. Figure 4.89 Transmission between a pair of identical 2 cm monopole antennas (25 ps/div).
90. Figure 4.90 Transmission between a pair of identical 2 cm monopole antennas (100 ps/div).
91. Figure 4.91 The designed T‐pulse in the time and frequency domains. (a) The designed T‐pulse in the time domain. (b) The spectral magnitude of the T‐pulse in the frequency domain fitting the FCC Mask. The vertical axis is in dB and the red line represents the level of the FCC UWB mask. (c) The phase of the T‐pulse in the frequency domain. The vertical axis represents the phase.
92. Figure 4.92 The schematic of a TEM horn.
93. Figure 4.93 The bicone and the TEM horn.
94. Figure 4.94 The output current at the TEM horn placed 1 m away.
95. Figure 4.95 Comparison of the waveforms for the two scenarios.
96. Figure 4.96 Comparison of waveforms between the T‐pulse and the normalized and shifted output pulses when the antenna separation is 1 m.
97. Figure 4.97 The output pulse from the TEM horn placed 10 m away.
98. Figure 4.98 Comparison between the T‐pulse and the normalized and shifted output current when the antenna distance is 10 m.
99. Figure 4.99 The electromagnetic simulation setup, showing the dimensions, spacing, and loading of the transmitting and receiving antennas.
100. Figure 4.100 (a) The power transfer efficiency. (b) The voltage induced efficiency.
101. Figure 4.101 Water‐filling method derived from Shannon's theorem showing the power allocated at one frequency segment.
102. Figure 4.102 Water‐filling method derived from Shannon's theorem showing the power allocated at 25 frequency segments.
103. Figure 4.103 Water‐filling method derived from Shannon's theorem showing the power allocated at 44 frequency segments.
104. Figure 4.104 Power delivered per number of bandwidth segments used.
105. Figure 4.105 An example of Gabor’s distribution for M = 101. As M increases, power is allocated over all bandwidth segments.
106. Figure 4.106 The capacity as a function of power delivered for P_N = 10⁻⁷ W.
107. Figure 4.107 The capacity as a function of the power delivered for P_N = 10⁻⁷ W with capacities based on a minimum SNR.
108. Figure 4.108 The capacity as a function of the power delivered for P_N = 10⁻⁷ W. Note that the SNR always exceeded 0 dB in this case.
109. Figure 4.109 A comparison of the Hartley, Nyquist, and Tuller capacity with the Shannon Capacity as a function of SNR for M = 20.
110. Figure 4.110 The capacity as a function of power delivered for P_N = 10⁻⁸ W.
111. Figure 4.111 The capacity as a function of power delivered for P_N = 10⁻⁹ W.
112. Figure 4.112 Antenna configuration considered in this section for simultaneous information and power transfer.
113. Figure 4.113 VSWR of the dipole antenna with different matching schemes.
114. Figure 4.114 Power efficiency of the dipole antenna with different matching schemes.
115. Figure 4.115 Voltage efficiency of the dipole antenna with different matching schemes.
116. Figure 4.116 Shannon capacity versus power delivery in the case without any impedance matching.
117. Figure 4.117 Flowchart of SRFT for broadband matching.
118. Figure 4.118 Matching network designed with SRFT.
119. Figure 4.119 Non‐Foster matching network.
120. Figure 4.120 Shannon capacity versus power delivery with a constraint of VSWR < 2.
121. Figure 4.121 Gabor capacity versus power delivery with a constraint of VSWR < 2.
122. Figure 4.122 HNT capacity versus power delivery with a constraint of VSWR < 2.
123. Figure 4.123 Shannon capacity versus power delivery with a constraint of VSWR < 3.
124. Figure 4.124 Gabor capacity versus power delivery with a constraint of VSWR < 3.
125. Figure 4.125 HNT capacity versus power delivery with a constraint of VSWR < 3.
126. Figure 4.126 Shannon capacity versus power delivery without constraint on VSWR.
127. Figure 4.127 Gabor capacity versus power delivery without constraint on VSWR.
128. Figure 4.128 HNT capacity versus power delivery without constraint on VSWR.
129. Figure 4.129 Configuration of a printed broadband matched dipole.
130. Figure 4.130 Fabricated printed broadband matched dipole.
131. Figure 4.131 VSWR of the printed broadband matched dipole.